1. Field of the Invention
The present invention relates to transitioning between different modulation modes in switching power converters (also referred to as switched mode power supplies) to reduce output voltage ripple and overshoot and undershoot during such transition.
2. Description of the Related Arts
In switched mode power supplies, power loss results from either current conduction loss or switching loss in the power switch. Under heavy load conditions, current is high and thus conduction loss is the dominating factor. However, under light load conditions, current is low and thus switching loss is the dominating factor.
To improve the light load efficiencies of switching power converters, multiple modulation modes are used to control the switching power converter at different output current (load) conditions. For example, in order to optimize the power efficiency for all load conditions, a switching power converter typically uses PWM (pulse width modulation) in heavy load conditions and PFM (pulse frequency modulation) in light load conditions.
In PWM mode, the switching power converter is controlled with a constant switching frequency, and therefore a constant switching period, but varies the duty cycle of the power switch in the switching power converter. Duty cycle refers to the fraction (often expressed as a percentage) of the switching period during which the power switch is ON. For example, a PWM switching scheme may have a switching frequency of 100 kHz and therefore a switching period of 10 μs. Hence, for a duty cycle of 30%, the power switch would be ON for 3 μs and OFF for 7 μs of each switching period. Under PWM control, the switching power converter regulates the output voltage based on feedback signals by adjusting the duty cycle of the power switch, but maintains a constant switching frequency.
In PFM mode, the switching power converter is controlled with the power switch being turned on with pulses of a set duration, but the duty cycle of the power switch is controlled with a variable switching frequency, and therefore a variable switching period. For example, a PFM switching scheme may turn on the power switch for 5 μs of each switching period, but vary the switching frequency between 40 kHz and 130 kHz. A switching frequency of 40 kHz would correspond to a switching period of 25 μs and therefore a duty cycle of 20%, whereas a switching frequency of 130 kHz would correspond to a switching period of 7.7 μs and therefore a duty cycle of 65%. Hence, under PFM control, the switching power converter regulates the output voltage based on feedback signals by adjusting the frequency and period of the power switch, but the power switch is ON for the same duration or for the durations corresponding to the same voltage-second product during each switching period.
FIG. 1A illustrates a conventional control scheme for a switching power converter. The switching power converter operates with two operating modes, PWM and PFM. When the output current (i.e., load) of the power converter is larger than load level (output current level) L0, the power converter operates in PWM mode with a constant switching frequency. However, when the load is smaller than load level L0, the power converter operates in PFM mode with the switching frequency decreasing as the load decreases.
With more than one modulation modes combined in the power converter control scheme, there exist transition points between the different operating modes. Note that the transition between PWM and PFM modes in the conventional control scheme of FIG. 1A is continuous. At load point L0, the power converter is at the transition point between the PWM and PFM modes, and can operate in PWM mode, PFM mode, or run back and forth between the PWM and PFM modes. If the power converter runs back and forth between PWM and PFM modes, the output voltage ripple typically becomes high. This is because PWM and PFM modes have to respond to the same control voltage while modulating different variables: pulse width and period. A perfect transition between the PWM and PFM modes requires that not only the control voltage requirements from the two operating modes are identical at the transition point 15, but also that the slope of the change in control voltage responding to the change in load is identical, which imposes a tough requirement for power converter design. Any discrepancy will cause the output voltage ripple to be higher than a normal, desired level.
FIG. 1B illustrates another conventional control scheme for a switching power converter. In this control scheme, a time lag is introduced for transitions between operating modes. That is, once the power converter enters an operation mode, it has to wait for the control loop to settle down before exiting that operation mode. In addition, control voltage hysteresis is introduced to minimize transition between operation modes. That is, the control voltage has to go beyond a level that represents a defined hysteresis Lhys in the load (output current) in order to transition into the other operating mode. For example, as shown in FIG. 1B, the load of a switching power converter in PWM mode would have to drop beyond load L0−Lhys to transition 20 to PFM mode, and the load of a switching power converter in PFM mode would have to increase beyond L0+Lhys to transition 10 to PWM mode. As a result, output voltage ripple caused by transition between operation modes can be reduced.
By introducing hysteresis, if the load does not deviate out of the hysteresis range, the power supply can operate stably in one modulation mode. However, if the hysteresis range is large, output voltage overshoot or undershoot may appear during the transition between operation modes, because the hysteresis may force the control voltage in one operation mode to go higher or lower than the control voltage in the other operation mode, resulting in a step function of the control voltage after the transition to the new operating mode. On the other hand, if the hysteresis range is too small, it may not be enough to prevent oscillation between operating modes. As a result, output voltage ripple may be higher due to not only imperfect slope matching, but also the hysteresis itself.
Another disadvantage of the control of FIGS. 1A and 1B is the wide control voltage range. For example, in a Flyback type switching power converter, the output power can be expressed as
            P      out        =                                        (                                          V                                  i                  ⁢                                                                          ⁢                  n                                            ⁢                              T                on                                      )                    2                          2          ⁢                      L            m                    ⁢                      T            p                              ⁢      η        ,where Pout is the output power, Vin is the rectified input voltage, Ton is the turn-on time of a power MOSFET switch, Lm is the magnetizing inductance of the transformer, Tp is the switching period, η is the conversion efficiency. If using VinTon control in PWM mode, the control variable is VinTon. If PWM control covers the load from 10% to 100%, the load ratio is 100%:10%=10:1. The control voltage ratio can be derived as √{square root over (10)}:1, or 3.16:1. In PFM mode, the control variable is Tp. If PFM covers the load range 10% to 0.5%, the control voltage ratio is 50:1. Thus, a much larger control voltage ratio needs to be supported by the PFM control. The wider control voltage range requirement in PFM mode may cause difficulty in implementation, particularly by the limitation that the highest control voltage in PFM mode cannot exceed the lowest control voltage in PWM mode.